A radio frequency amplifier has at least two principal functions. It must first amplify an input RF signal by an accurately known gain factor to produce a desired output level. Secondly, the input and output RF signals must be efficiently coupled into and out of the amplifier at the desired RF frequency (often over a broad band of frequencies) with little or no mismatch in RF impedances. For example, the output amplifier stage must efficiently couple an amplified r.f. signal to an RF load connected to its output terminals. The output stage must be capable of delivering the desired amount of RF signal power to the load with acceptably low levels of signal distortion if the amplifier is operated in a linear class (e.g., class A, class AB or class B). Moreover, the output stage should have a gain that is not greatly dependent upon load impedance. A well-designed RF output amplifier stage should achieve these performance specifications while consuming low quiescent power, providing stable operation under all expected input and output conditions, and without limiting the frequency response of the amplifier. Unfortunately, these (and perhaps other) well known desired performance specifications often can be difficult to simultaneously obtain from RF amplifier stages operated at very high frequencies (e.g., on the order of hundreds of MHz and in the GHz ranges).
In high-frequency RF communications equipment, considerable expense and circuitry has typically been necessary to design and construct intermediate amplifier stages which can develop sufficient power to drive high-power output stages. At high frequencies, stability problems become so severe that often per stage gains of only 5 to 8 dB are practical. The associated per stage band limiting components are generally expensive and consume precious space in modern RF communications equipment (miniaturization and minimization of which is usually desirable for a number of reasons).
Stability problems are primarily caused by uncertain amounts of reflected feedback impedance related to the effective capacitance of the active components in addition to layout-dependent stray capacitance. Furthermore, the feedback capacitance (commonly referred to as "Miller" capacitance) is one of the primary causes of changes in amplifier input impedance as load impedance varies with frequency.
Unavoidable capacitance between the collector and the base structures of a bipolar junction transistor (or between the source and the gate for a field-effect transistor) can cause a relatively large effective capacitance (the elemental component capacitance multiplied by 1 plus the voltage gain of the stage) to appear in parallel with the input to the transistor. The process by which the component capacitance is magnified and presented in parallel with the input terminals is often referred as the Miller effect. The Miller effect can reduce the unity-gain crossover frequency (gain-bandwidth product) of a transistor amplifier substantially if the amplifier is not well-designed. Because the amount of Miller feedback capacitance is related to stage gain, such feedback capacitance increases as the stage gain increases. In the case of a tuned load, this effective input capacitance typically can vary widely. In a production environment, these uncertain capacitances vary from one lot of amplifiers to another, often requiring expensive and critical manual tuning and/or manual "trimming" of passive components.
An additional need to conserve DC power exists whenever an amplifier stage is to be used in battery-operated equipment. Excess DC current is typically required so as to set the DC bias current of an RF amplifier stage at an appropriate level to insure a desired RF output power. (Such excess current is, for example, sometimes required to overcome biasing changes caused by expected transistor gain variations between different units in a large scale manufacturing environment.) Unfortunately, such excess current, when drawn from a battery power supply, can seriously shorten the life of the power supply, thereby reducing equipment reliability and increasing equipment maintenance costs.
FIG. 1 is a schematic diagram of one commonly-used prior art discrete high-frequency power amplifier 10. An input signal to is coupled to amplifier 10 via a coupling capacitor 12 and a matching network 14 (the purpose of which is to match the input impedance of the amplifier with the impedance of the signal source and thereby obtain more efficient signal transfer). The RF output of amplifier 10 is coupled from the collector of transistor 20 to a load 16 via another coupling capacitor 18. Amplifier 10 typically includes a bipolar junction transistor 20 connected in the common-emitter configuration, a radio frequency (RF) choke 22 (L1), bias voltage divider resistors 24 and 26, and an RF bypass capacitor 28. Supply voltage V.sub.cc is connected to the collector of transistor 20 via RF choke 22 which, together with bypass capacitor 28, prevent significant RF energy from reaching the power supply.
While the circuit configuration shown in FIG. 1 is extremely useful in a wide variety of different applications, it also has a lot of disadvantages--all of which become more critical as operating frequency increases and/or as the desired frequency band of operation is broadened. The input and output capacitance of amplifier 10 may be unacceptably high due to the Miller effect discussed above. Moreover, input and output capacitance may vary widely with amplifier instantaneous gain, load, signal level and frequency. There is a trade-off in amplifier 10 between stability and high gain which seriously limits the maximum power gain that can be expected. Moreover, implementation of amplifier 10 in an integrated circuit environment poses still further problems, since wire bonding inductance can become critical at high power levels and result in higher packaging costs. An integrated circuit implementation of amplifier 10 typically also requires the back surface of the integrated circuit chip to be tied to the output terminal, and thus requires ground isolation techniques to be used in the "packaging" of the IC chip.
Other high-frequency amplifier circuits exist which are better adapted for implementation in an integrated circuit format. For instance, U.S. Pat. No. 4,240,041 to Hashimoto et al (1980) discloses a high-frequency amplifier circuit capable of being integrated onto a single semiconductor chip. The output transistor of the amplifier disclosed may be operated in class AB, B or C by setting circuit parameters as desired. Moreover, the amplifier disclosed in this reference can be placed on a chip having relatively few (5) leads. However, the disclosed amplifier circuit uses double inversion (i.e., the output voltage is of the same polarity as the input voltage), thereby creating the possibility that positive feedback from the output to the input terminal may cause instability at high frequencies. Moreover, the amplifier gain is a complex function of the biasing resistance and the relative areas and transconductances of the various transistors used in the circuit. Finally, the biasing resistance must be relatively small to mask expected variations in transistor gains, thus causing power to be wasted.
Other examples of prior art possibly relevant to the present invention include (and there may be many more):
U.S. Pat. No. 3,392,342 to Ordower--(1968); PA0 U.S. Pat. No. 3,626,313 to Zuk--(1971); PA0 U.S. Pat. No. Re. 30,297 to Wittlinger--(1980); PA0 U.S. Pat. No. 3,992,676 to Knight--(1976); PA0 U.S. Pat. No. 3,942,129 to Hall--(1976); PA0 U.S. Pat. No. 3,950,708 to Hall--(1976); PA0 U.S. Pat. No. 3,952,257 to Schade, Jr.--(1976); PA0 U.S. Pat. No. 4,028,631 to Ahmed--(1977); PA0 U.S. Pat. No. 4,140,977 to Ahmed--(1979); PA0 U.S. Pat. No. 4,237,414 to Stein--(1980); and PA0 U.S. Pat. No. 4,242,643 to Leidich--(1980).
The patents listed above disclose various bipolar junction transistor configurations which can be used in high-frequency amplifier applications. For instance, U.S. Pat. No. 3,392,342 to Ordower discloses current mirror biasing. Zuk teaches a current mirror amplifier employing scaled geometries for the input and output transistors and using a cascaded output arrangement. Wittlinger teaches a current mirror amplifier which maintains collector potentials of the input and output transistors at equal levels. Knight shows a current mirror amplifier similar to Wittlinger's with a self-biased cascaded output stage. Ahmed '631 teaches a current mirror amplifier having a collector-base shunted input biasing resistor used to reduce input impedance.
Despite the recent advantages in integrated circuit technology, and despite the existence of a multitude of prior art circuits such as those identified above, no monolithic (i.e. integrated circuit) RF amplifier building blocks previously have been developed which can provide truly accurate and relatively high gain and power levels over a very wide band of high RF frequencies while yet permitting accurate control of DC output current for a given DC input current and minimizing the DC input power for a given RF output power. Moreover, most integrated-circuit high-frequency amplifier stages which have been proposed have significant high-frequency stability problems due to excessive input capacitance which changes over a large range with changing load impedance. Critical factory trim of individual components due to capacitance variations in active amplifier elements typically has been necessary, in the past, to obtain desired gain factors, and cascaded stages are typically needed to drive RF power devices or modules even when such trimming techniques are employed. No integrated circuit building block has previously been available which solves all of these problems at low cost and low complexity. Such a building block, if available, could be used in a variety of RF communications equipment product lines including mobile, cellular and hand-held battery communications equipment. If such a building block were capable of achieving stability at high efficiency, it would be ideal for use in a wide variety of different applications in such equipment, such as for RF preamplification, RF power drivers, RF power modules, and receiver mixers or IF amplifiers.